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| HV9921/22 Initial Release 3-Pin Switch-Mode LED Lamp Driver IC Features Constant Output Current: o HV9921 - 20mA o HV9922 - 50mA Universal 85-264VAC Operation Fixed OFF-Time Buck Converter Internal 500V Power MOSFET General Description The HV9921/22 are pulse width modulated (PWM) high-efficiency LED driver control ICs. They allow efficient operation of LED strings from voltage sources ranging up to 400VDC. The HV9921/22 include an internal high-voltage switching MOSFET controlled with fixed off-time TOFF of approximately 10s. The LED string is driven at constant current, thus providing constant light output and enhanced reliability. The output current is internally fixed at 20mA for HV9921 and 50mA for HV9922. The peak current control scheme provides good regulation of the output current throughout the universal AC line voltage range of 85 to 264VAC or DC input voltage of 20 to 400V. Applications Decorative Lighting Low Power Lighting Fixtures Typical Application Circuit ~ AC ~ CIN LED1 BR1 C0 LEDN D1 L1 3 VDD DRAIN 1 U1 CDD HV9921/22 GND 2 NR042205 NR030805 HV9921/22 Ordering Information Package Options DEVICE TO-92 HV9921 HV9922 HV9921N3 HV9922N3 SOT-89 HV9921N8 HV9922N8 Absolute Maximum Ratings Supply Voltage, VDD Supply Current, IDD Operating Ambient Temperature Range Operating Junction Temperature Range Storage Temperature Range Power Dissipation @ 25C, TO-92 Power Dissipation @ 25C, SOT-89 -0.3 to +10V +5mA -40C to +85C -40 to +125C -65 to +150C 740mW 1600mW Mounted on FR4 board, 25mm x 25mm x 1.57mm. Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. (The * denotes the specifications which apply over the full operating junction temperature range of -40C < TA < +85C, otherwise the specifications are at TA=25C, VDRAIN=50V, unless otherwise noted) Electrical Characteristics Regulator (VDD) Symbol VDD VDRAIN VUVLO VUVLO IDD Parameter VDD Regulator Output VDRAIN Supply Voltage Min 7.1 20 5.0 Typ 7.5 Max 7.8 Units V V V Conditions VDD Undervoltage Threshold VDD Undervoltage Lockout Hysteresis Operating Supply Current 200 200 Min 500 Typ 110 100 1 100 150 350 Max 210 200 5 mV A Units V pF mA VDD(EXT) = 8.5V, VDRAIN = 40V Conditions IDRAIN = 20mA IDRAIN = 50mA VDRAIN = 400V Output (DRAIN) Symbol Parameter VBR Breakdown Voltage RON RON CDRAIN ISAT ON Resistance - HV9921 ON Resistance - HV9922 Output Capacitance MOSFET Saturation Current Current Sense Comparator Symbol Parameter ITH Threshold Current - HV9921 ITH TBLANK TON(MIN) Threshold Current - HV9922 Leading Edge Blanking Delay Minimum ON Time Min 20.5 52 200 300 Typ Max 25.5 63 400 650 Units mA * mA ns ns * * Conditions OFF-Time Generator Symbol Parameter TOFF OFF Time Min 8 Typ 10.5 2 Max 13 Units s Conditions NR040805 HV9921/22 Functional Block Diagram GND VDD DRAIN TOFF = 10s Regulator 7.0V REF S + R Q Q R TBLANK = 300ns HV9921/22 Pin Configuration DRAIN - This is a drain terminal of the output switching MOSFET and a linear regulator input. VDD - This is a power supply pin for all control circuits. Bypass this pin with a 0.1uF low impedance capacitor. GND - This is a common connection for all circuits. TO-243AA (SOT-89) TO-92 3 NR040805 HV9921/22 Typical Performance Characteristics (T 25 J = 25C unless otherwise noted) 200 HV9921 24 Threshold Current, mA ON Resistance, Ohm 180 160 140 120 100 80 60 40 20 0 -40 23 22 21 20 19 -40 -15 10 35 60 85 110 -15 10 35 60 85 110 Junction Tem perature, C Junction Tem perature, C 12 10 OFF Time, uS 8 6 4 2 0 -40 DRAIN Capacitance (pF) 1000 100 10 1 -15 10 35 60 85 110 0 10 20 DRAIN Voltage (V) 30 40 Junction Tem perature, C 580 DRAIN Breakdown Voltage, V 570 560 550 540 530 520 510 500 490 -40 -15 10 35 60 85 110 180 160 DRAIN Current, mA 140 120 100 80 60 40 20 0 0 10 20 DRAIN Voltage, V 30 40 TJ = 25C TJ = 125C Junction Tem perature, C 4 NR040805 HV9921/22 Functional Description HV9921 and HV9922 are PWM peak current controllers for controlling a buck converter topology in continuous conduction mode (CCM). The output current is internally preset at 20mA (HV9921) or 50mA (HV9922). When the input voltage of 20 to 400V appears at the DRAIN pin, the internal high-voltage linear regulator seeks to maintain a voltage of 7VDC at the VDD pin. Until this voltage exceeds the internally programmed under-voltage threshold, the output switching MOSFET is non-conductive. When the threshold is exceeded, the MOSFET turns on. The input current begins to flow into the DRAIN pin. Hysteresis is provided in the under-voltage comparator to prevent oscillation. When the input current exceeds the internal preset level, a current sense comparator resets an RS flipflop, and the MOSFET turns off. At the same time, a one-shot circuit is activated that determines the duration of the off-state (10.5S typ.). As soon as this time is over, the flip-flop sets again. The new switching cycle begins. A "blanking" delay of 300nS is provided that prevents false triggering of the current sense comparator due to the leading edge spike caused by circuit parasitics. VO is the forward voltage of the LED string. TOFF is the off-time of the HV9921/22. The output current in the LED string (IO) is calculated then as: I O = I TH - 1 I O , 2 (2) where ITH is the current sense comparator threshold. The ripple current introduces a peak-to-average error in the output current setting that needs to be accounted for. Due to the constant off-time control technique used in the HV9921/22, the ripple current is independent of the input AC or DC line voltage variation. Therefore, the output current will remain unaffected by the varying input voltage. Adding a filter capacitor across the LED string can reduce the output current ripple even further, thus permitting a reduced value of L1. However, one must keep in mind that the peak-to-average current error is affected by the variation of TOFF. Therefore, the initial output current accuracy might be sacrificed at large ripple current in L1. Another important aspect of designing an LED driver with HV9921/22 is related to certain parasitic elements of the circuit, including distributed coil capacitance of L1, junction capacitance and reverse recovery of the rectifier diode D1, capacitance of the printed circuit board traces CPCB and output capacitance CDRAIN of the controller itself. These parasitic elements affect the efficiency of the switching converter and could potentially cause false triggering of the current sense comparator if not properly managed. Minimizing these parasitics is essential for efficient and reliable operation of HV9921/22. Coil capacitance of inductors is typically provided in the manufacturer's data books either directly or in terms of the self-resonant frequency (SRF). SRF = 1 /(2 L C L ) , Application Information The HV9921/22 is a low-cost off-line buck converter IC specifically designed for driving multi-LED strings. It can be operated from either universal AC line range of 85 to 264VAC, or 20 to 400VDC, and drives up to tens of high brightness LEDs. All LEDs can be run in series, and the HV9921/22 regulates at constant current, yielding uniform illumination. The HV9921/22 is compatible with triac dimmers. The output current is internally fixed at 20mA for HV9921 and 50mA for HV9922. Both parts are available in space saving TO-92 and SOT-89 packages. Selecting L1 and D1 There is a certain trade-off to be considered between optimal sizing of the output inductor L1 and the tolerated output current ripple. The required value of L1 is inversely proportional to the ripple current IO in it. L1 = VO TOFF I O where L is the inductance value, and CL is the coil capacitance.) Charging and discharging this capacitance every switching cycle causes highcurrent spikes in the LED string. Therefore, connecting a small capacitor CO (~10nF) is recommended to bypass these spikes. Using an ultra-fast rectifier diode for D1 is recommended to achieve high efficiency and reduce the risk of false triggering of the current sense comparator. Using diodes with shorter reverse recovery time trr and lower junction capacitance CJ achieves better performance. The reverse voltage 5 NR040805 (1) HV9921/22 rating VR of the diode must be greater than the maximum input voltage of the LED lamp. The total parasitic capacitance present at the DRAIN pin of the HV9921/22 can be calculated as: PSWITCH 1 (V AC C P + 2 I SAT t rr )(V AC - -1 VO 2 TOFF ) (8) VAC is the input AC line voltage. The switching power loss associated with turn-off transitions of the DRAIN pin can be disregarded. Due to the large amount of parasitic capacitance connected to this switching node, the turn-off transition occurs essentially at zero-voltage. Conduction power loss in the HV9921/22 can be calculated as PCOND = D I O RON + I DD VIN (1 - D ) , 2 C P = C DRAIN + C PCB + C L + C J (3) When the switching MOSFET turns on, the capacitance CP is discharged into the DRAIN pin of the IC. The discharge current is limited to about 150mA typically. However, it may become lower at increased junction temperature. The duration of the leading edge current spike can be estimated as: TSPIKE V C = IN P + t rr I SAT (9) (4) where D = VO /VIN is the duty ratio, RON is the ON resistance, IDD is the internal linear regulator current. When the LED driver is powered from the full-wave rectified AC line input, the exact equation for calculating the conduction loss is more cumbersome. However, it can be estimated using the following equation: PCOND = K C I O RON + K d I DD V AC , 2 In order to avoid false triggering of the current sense comparator, CP must be minimized in accordance with the following expression: CP < I SAT (TBLANK ( MIN ) - t rr ) , V IN ( MAX ) (5) where TBLANK(MIN) is the minimum blanking time of 200ns, and VIN(MAX) is the maximum instantaneous input voltage. Estimating Power Loss Discharging the parasitic capacitance CP into the DRAIN pin of the HV9921/22 is responsible for the bulk of the switching power loss. It can be estimated using the following equation: V 2C PSWITCH = IN P + VIN I SAT t rr FS , 2 (10) where VAC is the input AC line voltage. The coefficients KC and Kd can be determined from the minimum duty ratio of the HV9921/22 0.7 0.6 (6) Kd( Dm) Kc( Dm) 0.5 0.4 where Fs is the switching frequency, ISAT is the saturated DRAIN current of the HV9921/22. The switching loss is the greatest at the maximum input voltage. The switching frequency is given by the following: FS = VIN - -1 VO VIN TOFF 0.3 0.2 (7) 0.1 0 0.1 0.2 0.3 Dm 0.4 0.5 0.6 0.7 where is the efficiency of the power converter. When the HV9921/22 LED driver is powered from the full-wave rectified AC input, the switching power loss can be estimated as: Fig. 1. Conduction Loss Coefficients KC and Kd EMI Filter As with all off-line converters, selecting an input filter is critical to obtaining good EMI. A switching side capacitor, albeit of small value, is necessary in order to ensure low impedance to the high frequency 6 NR040805 HV9921/22 switching currents of the converter. As a rule of thumb, this capacitor should be approximately 0.10.2 F/W of LED output power. A recommended input filter is shown in Figure 2 for the following design example. Design Example 1 Let us design an HV9921 LED lamp driver meeting the following specifications: Input: Universal AC, 85-264VAC Output Current: 20mA Load: String of 10 LED (LW541C by OSRAM VF = 4.1V max. each) Step 1. Calculating L1. The output voltage VO = 10 VF 41V (max.). Use equation (1) assuming a 30% peak-to-peak ripple. L1 = 41V 10.5s = 72mH 0.3 20mA Switching power loss: PSWITCH 1 (264V 31 pF + 2 100mA 20ns ) 264V - 41V 2 10.5s 0.7 PSWITCH 120mW Minimum duty ratio: Dm = 41V /(0.7 264V 2 ) 0.16 Conduction power loss: PCOND = 0.25 (20mA) 210 + 0.63 200A 264V 55mW 2 Total power dissipation in HV9921: PTOTAL = 120mW + 55mW = 175mW Step 6. Selecting input capacitor CIN Output Power = 41V 20mA = 820mW Select L1 68mH, I=30mA. Typical SRF=170KHz. Calculate the coil capacitance. CL = 1 1 = 13 pF 2 L1 ( 2 SRF ) 68mH ( 2 170 KHz ) 2 Select CIN ECQ-E4104KF by Panasonic (0.1F, 400V, Metalized Polyester Film). Design Example 2 Let us design an LED lamp driver using the HV9922 that would meet the following specifications: Input: Universal AC, 85-135VAC Output Current: 50mA Load: String of 12 LED (Power TOPLED(R) by OSRAM, VF = 2.5V max. each) Step 1. Calculating L1. The output voltage VO = 12 VF = 30V (max.). Use equation (1) assuming a 30% peak-to-peak ripple. L1 = 30V 10.5s = 21mH 0.3 50mA Step 2. Selecting D1 Usually, the reverse recovery characteristics of ultrafast rectifiers at IF=20~50mA are not provided in the manufacturer's data books. The designer may want to experiment with different diodes to achieve the best result. Select D1 MUR160 with VR = 600V, trr 20ns (IF=20mA, IRR=100mA) and CJ 8pF (VF>50V). Step 3. Calculating total parasitic capacitance using (3) C P = 5 pF + 5 pF + 13 pF + 8 pF = 31 pF Select L1 22mH, I=60mA. Typical SRF=270KHz. Calculate the coil capacitance. CL = 1 1 = 15 pF 2 L1 (2 SRF ) 22mH ( 2 270 KHz ) 2 Step 4. Calculating the leading edge spike duration using (4), (5) TSPIKE = 264V 2 31 pF + 20ns 136ns < TBLANK ( MIN ) 100mA Step 2. Selecting D1 Select D1 MUR160 with VR = 600V, trr 50ns and CJ 8pF (VF>50V). Step 5. Estimating power dissipation in HV9921 at 264VAC using (8) and (10) Let us assume that the overall efficiency = 0.7. 7 NR040805 HV9921/22 Step 3. Calculating total parasitic capacitance using (3) C P = 5 pF + 5 pF + 15 pF + 8 pF = 33 pF Minimum duty ratio: Dm = 30V /(0.7 135V 2 ) 0.23 Step 4. Calculating the leading edge spike duration using (4), (5) TSPIKE = 135V 2 33 pF + 50ns 113ns < TBLANK ( MIN ) 100mA Conduction power loss: PCOND = 0.32 (50mA) 200 + 0.62 200A 135V 175mW 2 Total power dissipation in HV9922: PTOTAL = 65mW + 175mW = 240mW Step 5. Estimating power dissipation in HV9922 at 135VAC using (8) and (10) Switching power loss: PSWITCH 1 (135V 33 pF + 2 100mA 50ns )135V - 30V 2 10.5s 0.7 Step 6. Selecting input capacitor CIN Output Power = 30V 50mA = 1.5W Select CIN 0.22F, 250V. PSWITCH 65mW Figure 2. Universal 85-264VAC LED Lamp Driver D2 D4 D3 CIN2 D5 LIN CIN CO D1 LED1 -LED12 AC Line 85-264V VRD1 F1 3 U1 L1 DRAIN 1 GND 2 HV9921 CDD VDD 8 NR040805 HV9921/22 Figure 3. Typical Efficiency 82.00 80.00 78.00 76.00 74.00 72.00 70.00 68.00 66.00 64.00 62.00 75 100 125 150 175 200 225 250 275 Input AC Line Voltage (VAC) Figure 4. Switch-Off Transition. Ch1: VDRAIN, Ch3: IDRAIN Efficiency (%) ZER O VO LT AG E T R A N S IT IO N Figure 5. Typical Efficiency Figure 6. Switch-Off Transition. Ch1: VDRAIN, Ch3: IDRAIN L E A D IN G E D G E S P IK E S W IT C H O F F 25 m A 9 NR042205 HV9921/22 HV9921/22 Layout Considerations See Figure 7 for a recommended circuit board layout for the HV9921/22. Single Point Grounding Use a single point ground connection from the input filter capacitor to the area of copper connected to the GND pin. Bypass Capacitor (CDD) The VDD pin bypass capacitor CDD should be located as near as possible to the VDD and GND pins. Switching Loop Areas The area of the switching loop connecting the input filter capacitor CIN, the diode D1 and the HV9921/22 together should be kept as small as possible. The switching loop area connecting the output filter capacitor CO, the inductor L1 and the diode D1 together should be kept as small as possible. Thermal Considerations vs. Radiated EMI The copper area where GND pin is connected acts not only as a single point ground, but also as a heat sink. This area should be maximized for good heat sinking, especially when HV9921N8 or HV9922N8 (SOT-89 package) are used. The same applies to the cathode of the free-wheeling diode D1. Both nodes are quiet and therefore, will not cause radiated RF emission. The switching node copper area connected to the DRAIN pin of the HV9921/22, the anode of D1 and the inductor L1 needs to be minimized. A large switching node area can increase high frequency radiated EMI. Input Filter Layout Considerations The input circuits of the EMI filter must not be placed in the direct proximity to the inductor L1 in order to avoid magnetic coupling of its leakage fields. This consideration is especially important when unshielded construction of L1 is used. When an axial input EMI filter inductor LIN is selected, it must be positioned orthogonal with respect to L1. The loop area formed by CIN2, LIN and CIN should be minimized. The input lead wires must be twisted together. Figure 7. Recommended circuit board layout with HV9921N3/HV9922N3 COMPONENT SIDE VIEW CO LED + D1 D2-5 CIN2 CIN CDD U1 L1 LED - VRD1 F1 AC Line 85-264VAC LIN Doc. # DSFP-HV9921/HV9922 NR042205 |
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